Field of the Invention
The present invention generally relates to digital radio communication apparatuses and, more particularly, to a digital mobile radio communication apparatus provided with an analog filter and a digital filter.
Recently, with the depletion of radio wave resources foreseen, the communication standards stipulate increasingly tight restriction of the use of a channel band width. Conventionally, such a restriction has been met by improving hardware elements and circuit technology. More specifically, the performance of an analog filter is improved for that purpose. As the requirements stipulated by the communication standards become more strict, it is demanded that a software approach be introduced to implement a digital filter or to complement the performance of an analog filter.
FIGS. 1-5 illustrate the technology used in a digital radio communication apparatus according to the related art.
FIG. 1A shows a model of a digital radio transmission system. Referring to FIG. 1A, T.sub.b (.omega.) indicates a low-pass filter characteristic of a transmission unit, T.sub.r (.omega.) indicates a band-pass filter characteristic, F.sub.r (.omega.) indicates a transfer characteristic of a transmission path (air), R.sub.r (.omega.) indicates a band-pass filter characteristic of a receiver, and R.sub.b (.omega.) indicates a low-pass filter characteristic of the receiver. An overall transfer characteristic H(.omega.) is given by
H(.omega.)=T.sub.b (.omega.)T.sub.rb (.omega.)F.sub.rb (.omega.)R.sub.rb (.omega.)R.sub.b (.omega.),
where T.sub.rb (.omega.) indicates an equivalent low-pass filter characteristic of T.sub.r (.omega.), F.sub.rb (.omega.) indicates an equivalent low-pass filter characteristic of F.sub.r (.omega.) and R.sub.rb (.omega.) indicates an equivalent low-pass filter characteristic of R.sub.r (.omega.)
When such a transmission system is to transmit a pulse signal G(.omega.) from a signal source, an input waveform for a discrimination circuit is given by EQU r(t)=1/2.pi..intg..sub.-.infin..sup..infin. G(.omega.)H(.omega.)e.sup.j.omega.t d.omega. (1)
FIG. 1B shows an eye pattern of an input waveform for the discrimination circuit. Assuming that the signal source transmits a .pi./4-shifted PSK modulated signal, there is no intersymbol interference occurring in the input waveform of the discrimination circuit if H(.omega.) satisfies the Nyquist condition. The eye aperture is open ((a) of FIG. 1B). However, if the Nyquist condition fails to be satisfied due to a variation of the performance of filter elements that has occurred in the process of fabrication, or due to a variation in the operating conditions (temperature, power-supply voltage, etc.), intersymbol interference occurs so that the eye aperture begins to close ((b) of FIG. 1B).
FIG. 1C shows a constellation (arrangement of codes) that illustrates the above-described relation. Generally, code points on the transmitting side ((a) of FIG. 1C) vary (are displaced) in the air as shown in (b) of FIG. 1C before arriving at the receiving side. If the combination of filters on the receiving side satisfies the Nyquist condition, the variation in the air settles to a state as shown in (c) of FIG. 1C at a discrimination point. That is, the intersymbol distance H at the discrimination point is relatively large. However, if there is a deviation in the filter characteristic on the reception side, intersymbol interference occurs so that it is impossible to properly restore the code points ((d) of FIG. 1C). That is, the intersymbol distance H at the discrimination point is relatively small.
FIG. 2 shows a relation between a cosine roll-off factor a and the constellation in the air. FIG. 2A shows the relation that occurs when .alpha.=0.8; FIG. 2B shows the relation that occurs when .alpha.=0.5; and FIG. 2C shows the relation that occurs when .alpha.=0.2. The smaller the factor .alpha., the smaller the occupied bandwidth so that the more preferable it is in terms of efficient use of the bandwidth. Accordingly, .alpha. tends to be controlled to maintain it at low level in current digital communication systems. However, the constellation in the air deviates from that of the point of origination as the level of .alpha. is lowered, requiring precise control of the receiver filter in order to restore the constellation.
Conventionally, in order to construct a receiver with a strict requirement for selectivity between adjacent channels, a high-performance analog filter formed of crystal or ceramic is used.
FIG. 3A shows a characteristic of attenuation of an analog filter with respect to frequency. Generally, in order to obtain a large attenuation, a plurality of analog filters are connected in multiple stages so as to produce a high performance (large attenuation). Such an approach causes the number of required elements to increase, and increases the size and cost of the resultant apparatus.
FIG. 3B shows a group delay characteristic of an analog filter with respect to frequency. The delay time of a signal varies with respect to the frequency. Therefore, connecting a plurality of analog filters to form multiple stages in an attempt to obtain a high-attenuation characteristic causes degradation in the group delay characteristic.
Further, a characteristic of analog elements is subject to a variation that occurs in the process of production. The characteristic also varies significantly with time and due to a variation in the operating conditions (temperature, power-supply voltage, etc.). Thus, it is difficult to implement and maintain the precise Nyquist characteristic.
According to one approach, an analog filter designed to eliminate out-of-band noise is used in the first stage, several stages of the receiving system are linearized, and the majority of the filter performance (the Nyquist characteristic, the attenuation characteristic, etc.) is implemented (covered) by the digital filter in a subsequent stage.
FIG. 4 shows a construction of a digital radio communication apparatus (portable terminal) according to the related art. The digital radio communication apparatus comprises an antenna 1; a transmission/reception branching switch 2 (C); a transmitter 3, a frequency synthesizer 4 (SYN), a receiver 5, including an RF amplifier (RFA) 6, a first mixer (x) 7, a second mixer 9 (x), analog band-pass filters (BPF) 8, 10, 12 formed of crystal or ceramic, IF amplifiers (IFA) 11, 13, a quadrature detecting unit (QDT) 14 using the QPSK system, an A/D converter (A/D) 15, adaptive transversal filters 16, 17 using a digital system, a discriminating circuit (DSC) 18, a clock generator (CG) 19, an automatic frequency controller (AFC) 20, a voltage controlled oscillator (VCO) 21, and an automatic gain controller (AGC) 25.
CG 19 generates (reproduces) a sampling clock signal SK and a data clock signal DK based on the edges of demodulated I/Q signals. AFC 20 detects frequency deflection of the IF signal based on the edges of the demodulated I/Q signals. An output of AFC 20 is input to DSC 18 and used in control of a discriminated phase (phase rotation by .pi./4-shifted QPSK and the like). The output of AFC 20 is input to VCO 21 and used to maintain the frequency of the IF signal at a regular level.
Further, the digital radio communication apparatus comprises a TDMA synchronization controller 31 for controlling timings according to the TDMA system; a codec (CODEC) 32 for converting a sound signal into codes; a baseband processor (BBP) 33 of the sound signal; a microphone (MIC) 34; a speaker (SPK) 35; a CPU 41 for performing main control (console control and call control including location registration, standby, call origination, call incoming, and handover) of the apparatus; a main memory (MM) 42 embodied by a RAM, a ROM and an EEPROM or the like for storing control programs executed by the CPU 42 and associated data; a console unit (CSL) 43 operated by a user, including a display unit 44 embodied by a liquid crystal or the like for displaying dial numbers and messages, and a keyboard (KBD) 45 provided with dial keys; and function keys, and a common bus 46 for the CPU 41.
The CPU 41 controls incoming and outgoing calls via the TDMA synchronization controller 31. In a call state, in which a call can proceed with respect to a destination terminal, the sound signal from the MIC 34 is sampled by the BBP 33 and converted thereby into PCM data. The CODEC 32 converts the output of the BBP 33 into code data. The TDMA synchronization controller 31 formats the output of the CODEC 32 to produce transmitted data TD. The transmitter 3 modulates the transmitted data TD into a .pi./4-shifted QPSK signal for transmission via the antenna 1.
The wave received by the antenna 1 is amplified by the RFA 6 and converted by the mixers 7 and 9 so as to produce a first IF signal and a second IF signal, respectively. IFAs 11, 13 and AGC 25 amplify the IF signals to have a predetermined level. The IF signals are subject to quadrature detection by ODT 14 to produce quadrature detection signals I and Q. The detection signals I and Q are subject to A/D conversion by A/D 15. ATFs 16 and 17 convert the signals I and Q into reproduced signals I and Q having minimum errors .epsilon..sub.i and .epsilon..sub.q, respectively, with respect to the code points. The reproduced signals I and Q are subject to discrimination by DSC 18 so as to produce received data RD. The received data RD is input to the TDMA synchronization controller 31 where code data of the sound is retrieved. The code data is converted into PCM data by the CODEC 32. The PCM data is converted into the sound signal and audibly output by SPK 35.
FIG. 5 shows a construction of an adaptive transversal filter according to the related art. The adaptive transversal filter comprises an adaptive transversal filter (ATF) 16/17, including a tap factor operator 16A, and a FIR (finite impulse response) filter 16B, and consisting of a delay circuit (Z.sup.-1) 16a, a multiplier (x) 16b, and an adder (.SIGMA.) 16c; and a discrimination unit (DSC) 18, including a discrimination circuit 18a for code points, and an error detection unit 18b.
An output y.sub.j of the FIR filter 16B is given by ##EQU1##
where a tap (weight) factor vector A.sub.j =[a.sub.0j, a.sub.ij, . . . , a.sub.Nj ].sup.T, and an input signal vector x.sub.j =[x.sub.j, x.sub.j-1, . . . , x.sub.j-N ].sup.T.
The discrimination circuit 18a compares the output y.sub.j with a code point d.sub.j so as to produce reproduced data RD closest to the code point d.sub.j. The error detection unit 18b compares the output y.sub.j with the code point d.sub.j so as to produce an error signal .epsilon..sub.j =d.sub.j -y.sub.j (=d.sub.j -A.sub.j.sup.T x.sub.j). The tap factor operator 16A obtains an optimum tap factor vector A.sub.j+1 =[a.sub.0j+1, a.sub.1j+1, . . . , a.sub.Nj+1 ].sup.T which causes the square of the error .epsilon..sub.j.sup.2 to have a minimum value.
The optimum tap factor vector A.sub.j+1 is obtained at the next instant using the weight vector method of Wiener. However, this method requires complex, large-volume operations to be carried out so that real-time processing, by a DSP or the like, is impossible when the number of taps N is increased. Accordingly, the LMS (least mean square) method is generally used to obtain a step-by-step approximation of the optimum tap factor vector A.sub.j+1. The LMS method is also called the steepest descent method. The tap factor vector A.sub.j+1 for the next instant is given by EQU A.sub.j+1 =A.sub.j -.mu..gradient..sub.j
where .mu. indicates a parameter for controlling a convergence speed/stability, and .gradient..sub.j indicates an instantaneous gradient.
The instantaneous gradient .gradient..sub.j is given by ##EQU2##
Accordingly, the following relation holds. EQU A.sub.j+1 =A.sub.j +2.mu..epsilon..sub.j X.sub.j
where the parameter .mu. is appropriately set. When .epsilon..sub.j =0, A.sub.j+1 =A.sub.j indicates an optimum tap factor vector.
A combination of the analog filter and the adaptive transversal filter as described above can be adapted for variations of the transmission path characteristic H(.omega.).
However, if the adaptive transversal filter is used, it is necessary to obtain a next-instant tap factor vector A.sub.j+1 for each symbol received, thus imposing a heavy load on the tap factor operator 16A. While the number of taps N need to be large in order to obtain a high-attenuation characteristic using the digital filter, the processing speed of a DSP or the like presents a bottleneck.
When the LMS method is used, the adaptive process starting with an initial vector A.sub.0 is such that, if the level of .mu. is low, the adaptive process proceeds with substantially no oscillation so that the optimum factor to produce the minimum value of .epsilon..sub.j.sup.2 is obtained smoothly. However, the convergence speed is low. If, on the other hand, .mu. is high, each of the adaptive steps goes too "far", causing an oscillation before arriving at the point that produces the minimum value of .epsilon..sub.j.sup.2. In this case, while the convergence speed is high, there is a likelihood that divergence may take place. That is, if the adaptive transversal filter is used, the receiving system might be instable.
The adaptive transversal filter is designed to minimize an error power .epsilon..sub.j.sup.2 with respect to the code point. With the adaptive transversal filter, it is impossible to know which of the characteristics of the filter of the receiving system (roll-off characteristic, attenuation characteristic, group delay characteristic, phase characteristic, etc.) is improved. In other words, it is impossible to compensate and control a specific characteristic of the filter of the receiving system.